This invention relates to a code division multiple access signal receiving apparatus and, more particularly, to a code division multiple access signal receiving apparatus, which is for receiving a spread-spectrum modulated signal and reproducing transmit data, in a code division multiple access communication system.
In a CDMA (Code Division Multiple Access) mobile communications system, a base station spread-spectrum modulates control information and user information of each user by employing different spreading code sequences, multiplexes the modulated information and transmits the same. Each mobile station in the system sends and receives information upon spreading and despreading the information using a spreading code sequence specified by the base station.
FIG. 16 is a block diagram of a CDMA transmitter in a base station that encodes, multiplexes and transmits the transmit data of a control channel and of a plurality of user channels. In the Figure, numerals 111 to 11n denote spread-spectrum modulators of respective control and user channels, each having a frame generator 21, a serial/parallel converter (S/P converter) 22 for converting frame data to parallel data, and a spreading circuit 23.
The frame generator 21 includes a transmission data generator 21a for generating serial transmit data D1, a control-data generator 21b for generating control data CNDT such as a pilot, and a framing circuit 21c for forming the serial data D1 into a block every prescribed number of bits and inserting the control data CNDT before and after every block to thereby form frames. The pilot signal allows a receiver to recognize the amount of phase rotation caused by transmission so that the data may be subjected to a phase rotation by an equivalent amount in the opposite direction.
The S/P converter 22 alternately distributes the frame data (the control data and transmit data) one bit at a time to convert the frame data to two sequences DI, DQ, namely I-component (in-phase component) data and Q-component (quadrature-component) data. The spreading circuit 23 includes a pn sequence generator 23a for generating a noise code (pn sequence) specific to the base station, a channel code generator 23b for generating a channel code specific to the control channel or user channel, an EXOR circuit 23c for outputting a spreading code C1 by taking the EOR (exclusive-OR) between the noise code and the channel code, and EXOR circuits 23d, 23e for performing spread-spectrum modulation by taking the exclusive-ORs between the data DI and DQ (symbols) of the two sequences, and the spreading code Cl.
Reference characters 12i denote a combiner for outputting an I-component code-multiplexed signal ΣVI by combining the I-component spread-spectrum modulated signals V1 output by the spread-spectrum modulators 111˜11n of the user channels; 12q a combiner for outputting a Q-component code-multiplexed signal ΣVQ by combining the Q-component spread-spectrum modulated signals VQ output by the spread-spectrum modulators 111˜11n; 13i, 13q FIR-type chip shaping filters for limiting the bands of the code-multiplexed signals ΣVI, ΣVQ; 14i, 14q DA converters for DA-converting the outputs of the filters 13i, 13q; 15 a quadrature modulator for applying QPSK quadrature modulation to the code-multiplexed signals ΣVI, ΣVQ of the I and Q components and outputting the modulated signal; 16 a transmit circuit for converting the frequency of the output signal of the quadrature modulator to a radio frequency, amplifying the high frequency and transmitting the result; and 17 an antenna.
FIG. 17 is a block diagram of a CDMA receiver in a mobile station. A radio unit 31 converts a high-frequency signal received by the antenna 30 to baseband signals by applying a frequency conversion (RF→IF conversion). An orthogonality detector 32 detects orthogonality of the baseband signals and outputs in-phase component (I-phase component) data and quadrature component (Q-component) data. In the orthogonality detector 32, reference characters 32a denote a receive-carrier generator; 32b a phase shifter for shifting the phase of the receive carrier by π/2; and 32c, 32d multipliers for multiplying the baseband signals by the receive carrier and outputting the I-component signal and the Q-component signal. Low-pass filters (LPF) 33a, 33b limit the bands of these output signals and AD converters 35a, 35b convert the I-and Q-component signals to digital signals and input the digital signals to a despreading circuit 41.
The despreading circuit 41 subjects the input I- and Q-component signals to despread processing using a code identical with the spreading code on the transmit side and outputs a reference signal (pilot signal) and an information signal. A phase compensator (channel estimation unit) 42 averages the voltages of the I- and Q-components of the pilot signal over a prescribed number of slots and outputs channel estimation signals It, Qt. A synchronous detector 43 restores the phases of despread information signals I′, Q′ to the original phases based upon a phase difference θ between a pilot signal contained in a receive signal and an already existing pilot signal. More specifically, the channel estimation signals It, Qt are cosine and sine components of the phase difference θ, and therefore the synchronous detector 43 demodulates the receive information signal (I,Q) (performs synchronous detection) by applying phase rotation processing to the receive information signal (I′,Q′) in accordance with the following equation using the channel estimation signal (It,Qt):       (                            I                                      Q                      )    =            (                                    It                                Qt                                                              -              Qt                                            It                              )        ⁢          (                                                  I              ′                                                                          Q              ′                                          )      
An error correction decoder 44 decodes the original transmit data by using the signal that enters from the synchronous detector 43 and outputs the decoded data.
In the above-described mobile wireless communication system, a base station usually cannot use a fixed directivity pattern for communication with mobile stations; it communicates using a non-directional antenna. However, not only is transmission by a non-directional antenna poor in power efficiency because radio waves are emanated also in directions in which a targeted mobile station is not present, but such transmission also degrades communication quality by subjecting mobile stations other than the targeted mobile station to interference. For this reason, practice has been to equally divide the 360° circumference of the base station so as to split the cell into a plurality of sectors (sector-shaped zones), and use a directional antenna in each sector, thereby mitigating interference.
FIG. 18 is a schematic structural view of a transceiver in a code division multiple access communication system for a case where a cell has been divided into sectors, and FIG. 19 is a flowchart of transceive processing. These illustrate an example of a case where user data signals Data1, Data2 are transmitted from a single base station BS to mobile stations MS1, MS2 in two sectors neighboring each other. Sectors Sec1, Sec2 in the base station BS have transceive antennas ANT1, ANT2 possessing separate directivities and, by virtue of the antenna directivities, take charge of the sending and receiving of signals to and from coverage areas (sectors) that are geographically independent of each other.
The user data signals Data1, Data2 undergo encoding processing, for error correction or the like, in channel coders CH-cod1, CH-cod2 in respective ones of the sectors Sec1, Sec2, and the processed signals are input to spreading circuits SC1, SC2. Encoded data Cdata1, Cdata2 is spread-spectrum modulated in the spreading circuits SC1, SC2 by mutually different spreading-code sequences PN1, PN2 generated by spreading-code generators PNG11, PNG12, whereby transmit signals Sdata1, Sdata2 are obtained. The spreading codes PN1, PN2 used in the sectors Sec1, Sec2 are produced from portions having different phases in an M sequence generated from the same generating polynomial. As a result, the spreading codes produced are such that the their mutual cross-correlation values take on small values on average.
In the receivers of the mobile stations MS1, MS2, on the other hand, despreading circuits RSC1, RSC2 apply despread demodulation to received signals Sdata1′, Sdata2′ using the same code sequences PN1′ (=PN1), PN2′ (=PN2), which are synchronized to the spreading-code sequences used by the base station. Channel decoders (CH-Dec1, CH-Dec2) decode receive signals Data1′, Data2′.
Depending upon geographical conditions in which the mobile stations MS1, MS2 find themselves, e.g., as a result of the mobile stations being present where the antenna directivities of the two sectors Sec1, Sec2 of the base station BS overlap each other, there are cases where the signals from the two antennas ATN1, ATN2 are received by antennas ANT1′, ANT2′ of the mobile stations simultaneously. In such cases, receive signals other than the desired receive signals received by the mobile stations become interference noise because the two transmit signals were spread-spectrum modulated by spreading-code sequences that differ from each other. This noise remains in the receive demodulated signals. However, since this interference noise when averaged takes on a small value decided by the cross-correlation value between the spreading codes, the effects thereof are eliminated by the error correction functions, etc., of the channel decoders CH-Dec1, CH-Dec2, and data identical with the user data transmitted by the transmitting side is reproduced as receiver outputs Data1′, Data2′.
Even if the cross-correlation characteristic has been selected so as to have a problem-free characteristic when averaged, there are instances where the cross-correlation characteristic deteriorates momentarily, as at the time of a high-speed data transmission that requires communication to be performed at a low spreading rate [=(symbol period)/(chip period)]. FIG. 20 is a diagram useful in describing a partial cross-correlation characteristic between pseudo-noise sequences (spreading-code sequences). This illustrates a case where the spreading rate that prevails when the two items of transmit data Data1, Data2 are spread-spectrum modulated is 4 (one symbol is spread by four chips). The cross-correlation value between the two spreading-code sequences (PN1, PN2) is expressed by how many chips of the same polarity exist at the same time per symbol (=four chip lengths). With regard to the symbol indicated by the shading in FIG. 20, three chips in one sequence have identical polarities with those of the other sequence over the duration of this one symbol. The cross-correlation value, therefore, is very large. If the cross-correlation characteristic thus is inferior locally and transmit data is spread-spectrum modulated and transmitted using this portion of the characteristic, mutual interference noise that prevails after despreading is performed on the receiving side becomes large in the above-mentioned symbol portion, as shown in FIG. 21 (see the shaded portions), and this invites local deterioration of the transmission characteristic. The major part of the local deterioration in the transmission characteristic is corrected by error-correction decoding processing in the channel decoders CH-Dec1, CH-Dec2 using signal-to-noise ratio information, and correct data is decoded as a result. In comparison with a case where there is no local deterioration, however, an increase in interference power of the entire system attendant upon an increase in average required power occurs in order to achieve a certain transmission quality. This brings about undesirable results, such as a decrease in system capacity (the number of users that can be accommodated)
Further, it is difficult to estimate the signal-to-noise ratio accurately on a per-symbol basis. It is also difficult, therefore, to execute receive processing properly by detecting an increase in local interference noise in a specific symbol contained in a receive signal.